Wireless communication device with a low noise receiver

ABSTRACT

A wireless communication device includes a radio frequency antenna and a transceiver. The transceiver includes a receiver having a switching architecture configured to generate a plurality of output phases within a local oscillator period based on the filtered RF signal and a respective plurality of local oscillator signals. The plurality of output phases can be organized into at least K groups where K is an integer of four or greater, and each nth group of the K groups includes nth and (n+K)th output phases of the plurality of output phases. The receiver can difference the nth and (n+K)th output phases of each respective group of the K groups, resulting in gain-added output phases.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a divisional of U.S. application Ser. No.14/718,817, filed May 21, 2015, which is a continuation of U.S. patentapplication Ser. No. 13/232,873, filed Sep. 14, 2011, which is acontinuation of International Application No. PCT/US2009/068212, filedDec. 16, 2009, which claims priority to and the benefit of the filingdate of U.S. Provisional Patent Application No. 61/160,858, filed Mar.17, 2009, the benefits of the filing dates of which are hereby claimedand the disclosures of which are incorporated herein by this reference.

BACKGROUND

Portable communication devices, such as cellular telephones, personaldigital assistants (PDAs), WIFI transceivers, and other communicationdevices transmit and receive communication signal at variousfrequencies. For efficient communication, the frequency of the transmitand receive signals is many times higher than the baseband informationsignal that carries the information to be communicated. Therefore, atransceiver must upconvert the transmit signal and downconvert thereceive signal.

Usually, one or more mixers are used to upconvert the transmit signaland downconvert the receive signal. In many radio frequency (RF)communication methodologies, and in a quadrature modulation methodologyin particular, a mixer can be implemented using a series of switchesthat switch differential components of a quadrature signal according toa local oscillator (LO) signal. The frequency of the LO signal is chosenso that a radio frequency signal mixed with the LO signal is convertedto a desired frequency.

Signal upconversion and signal downconversion is performed by usingmixers, which are typically implemented using semiconductor switches. Indeep sub-micron technology the availability of passive switchesproviding low noise operation and highly efficient operatingcharacteristics enables the use of passive mixers where low currentconsumption and high performance is desired. Rail to rail voltages usedin the switch clock path and issues due to poor isolation between thein-phase (I) and quadrature-phase (Q) paths in the mixer imposelimitations on the use of a passive mixer.

A surface acoustic wave (SAW) filter is typically used to protect thereceive frequency band from interfering signals that may be out of thereceive band, but that may still cause interference, particularly atcertain multiples (harmonics) of the receive frequency. An LNA istypically used to amplify the relatively weak receive signal so that theinformation contained therein can be extracted. For a multibandreceiver, a separate SAW filter is needed for each band, and a separateLNA is needed to accept the output of each SAW filter. Thus SAW filtersand LNAs typically add complexity to the receiver architecture. Further,the LNAs consume power, and this power consumption must be sufficientlyhigh to allow the LNAs to pass large blocking signals withoutcompressing small desired signals.

Therefore, it would be desirable to have a low noise receiverarchitecture that may not rely on these additional elements.

SUMMARY

Embodiments of a low noise receiver include a downconverter configuredto receive a radio frequency (RF) signal, the downconverter comprising aswitching architecture configured to generate a plurality of outputphases based on a respective plurality of local oscillator (LO) signals,a differencing circuit configured to combine the plurality of outputphases such that an nth output phase is differenced with an (n+K)thoutput phase, resulting in gain-added output phases, and a summationfilter configured to receive the gain-added output phases and configuredto combine the gain-added output phases such that a response of thereceiver effectively reduces odd harmonics of the RF signal.

Other embodiments are also provided. Other systems, methods, features,and advantages of the invention will be or become apparent to one withskill in the art upon examination of the following figures and detaileddescription. It is intended that all such additional systems, methods,features, and advantages be included within this description, be withinthe scope of the invention, and be protected by the accompanying claims.

BRIEF DESCRIPTION OF THE FIGURES

The invention can be better understood with reference to the followingfigures. The components within the figures are not necessarily to scale,emphasis instead being placed upon clearly illustrating the principlesof the invention. Moreover, in the figures, like reference numeralsdesignate corresponding parts throughout the different views.

FIG. 1 is a block diagram illustrating a simplified portabletransceiver.

FIG. 2 is a schematic diagram of an embodiment of a known single-endedvoltage-mode downconverter implemented as a passive mixer using anapproximate 25% duty cycle topology.

FIG. 3 is a graphical illustration showing the LO signals used in anembodiment of the passive mixer described in FIG. 2.

FIG. 4 is a schematic diagram illustrating an embodiment of a low noisereceiver.

FIG. 5 is a graphical illustration showing an example frequency spectrumwithin which the low noise receiver operates.

FIG. 6 is a schematic diagram illustrating an alternative embodiment ofthe low noise receiver of FIG. 4.

FIG. 7 is a schematic diagram illustrating another alternativeembodiment of the low noise receiver of FIG. 4.

FIG. 8 illustrates a method of generating a waveform in which 3^(rd) and5^(th) harmonics are rejected.

FIG. 9 is a graphical illustration showing the derivation of the eightLO phases utilized by the low noise receiver of FIG. 7 for the case ofk=4.

FIG. 10 is a graphical illustration showing the effective quadrature LOwaveforms, each with 3^(rd) and 5^(th) harmonics rejected, that aregenerated by weighted combining of the eight LO phases utilized by thelow noise receiver of FIG. 7 for the case of K=4.

FIG. 11 is a schematic diagram illustrating an embodiment of a low noisereceiver that implements the effective quadrature LO waveforms of FIG.10.

FIG. 12 is a graphical illustration showing an example of the frequencyresponse of embodiments of the low noise receiver.

FIG. 13 is a schematic diagram illustrating an alternative embodiment ofthe low noise receiver of FIG. 11.

FIGS. 14A through 14D are graphical illustrations showing an examplefrequency response of an embodiment of the low noise receiver of FIG. 4at a receive frequency of 1 GHz.

DETAILED DESCRIPTION

Although described with particular reference to a portable transceiver,the SAW-less, LNA-less low noise receiver (also referred to herein asthe low noise receiver), can be used in any device that uses signaldownconversion in a receiver.

For a quad-band communication device operating in the GSM/EDGE frequencyspectrum, the low noise receiver described herein eliminates fourexternal SAW filters and on-chip low noise amplifiers (LNAs) that aretypically used in quad-band cell phone solutions, leading to large costand area savings. The elimination of the SAW filters and LNAs isachieved, at least in part, by implementing the highly linear, lownoise, passive, mixer architecture mentioned above, and partly by thecareful design of input and output matching circuitry.

The low noise receiver can be implemented in hardware, or a combinationof hardware and software. When implemented in hardware, the passivemixer and high Q RF filter using a passive mixer can be implementedusing specialized hardware elements and logic. When the low noisereceiver is implemented partially in software, the software portion canbe used to precisely control the various components. The software can bestored in a memory and executed by a suitable instruction executionsystem (microprocessor). The hardware implementation of the low noisereceiver can include any or a combination of the following technologies,which are all well known in the art: discrete electronic components, adiscrete logic circuit(s) having logic gates for implementing logicfunctions upon data signals, an application specific integrated circuithaving appropriate logic gates, a programmable gate array(s) (PGA), afield programmable gate array (FPGA), etc.

The software for low noise receiver comprises an ordered listing ofexecutable instructions for implementing logical functions, and can beembodied in any computer-readable medium for use by or in connectionwith an instruction execution system, apparatus, or device, such as acomputer-based system, processor-containing system, or other system thatcan fetch the instructions from the instruction execution system,apparatus, or device and execute the instructions.

In the context of this document, a “computer-readable medium” can be anymeans that can contain, store, communicate, propagate, or transport theprogram for use by or in connection with the instruction executionsystem, apparatus, or device. The computer readable medium can be, forexample but not limited to, an electronic, magnetic, optical,electromagnetic, infrared, or semiconductor system, apparatus, device,or propagation medium. More specific examples (a non-exhaustive list) ofthe computer-readable medium would include the following: an electricalconnection (electronic) having one or more wires, a portable computerdiskette (magnetic), a random access memory (RAM), a read-only memory(ROM), an erasable programmable read-only memory (EPROM or Flash memory)(magnetic), an optical fiber (optical), and a portable compact discread-only memory (CDROM) (optical). Note that the computer-readablemedium could even be paper or another suitable medium upon which theprogram is printed, as the program can be electronically captured, viafor instance, optical scanning of the paper or other medium, thencompiled, interpreted or otherwise processed in a suitable manner ifnecessary, and then stored in a computer memory.

FIG. 1 is a block diagram illustrating a simplified portable transceiver100. Embodiments of the low noise receiver can be implemented in any RFreceiver, RF transmitter or RF transceiver, and in this example, areimplemented in an RF receiver 120 associated with a portable transceiver100. The portable transceiver 100 illustrated in FIG. 1 is intended tobe a simplified example and to illustrate one of many possibleapplications in which the low noise receiver can be implemented. Onehaving ordinary skill in the art will understand the operation of aportable transceiver. The portable transceiver 100 includes atransmitter 110, a receiver 120, a baseband subsystem 130, adigital-to-analog converter (DAC) 160 and an analog-to-digital converter(ADC) 170. The transmitter 110 includes a modulator 116 and anupconverter 117. In an embodiment, the upconverter 117 can be asubsystem of the modulator 116. In alternative embodiments, theupconverter 117 can be a separate circuit block or circuit element.

The transmitter also includes any other functional elements thatmodulate and upconvert a baseband signal. The receiver 120 includesfilter circuitry and a downconverter 200 that enable the recovery of theinformation signal from the received RF signal. The downconverter 200implements portions of and embodiments of the low noise receiver, asdescribed herein.

The portable transceiver 100 also includes a power amplifier 140. Theoutput of the transmitter 110 is provided over connection 112 to thepower amplifier 140. Depending on the communication methodology, theportable transceiver may also include a power amplifier control element(not shown).

The receiver 120 and the power amplifier 140 are connected to a frontend module 144. The front end module 144 can be a duplexer, a diplexer,or any element that separates the transmit signal from the receivesignal. The front end module 144 also contains appropriate bandswitching devices to control the application of a received signal to thereceiver 120. The front end module 144 is connected to an antenna 138over connection 142.

In transmit mode, the output of the power amplifier 140 is provided tothe front end module 144 over connection 114. In receive mode, the frontend module 144 provides a receive signal to the receiver 120 overconnection 146.

If portions of the low noise receiver are implemented in software, thenthe baseband subsystem 130 also includes receiver software 155 that canbe executed by a microprocessor 135, or by another processor, to controlat least some of the operation of the low noise receiver to be describedbelow.

When transmitting, the baseband transmit signal is provided from thebaseband subsystem 130 over connection 132 to the DAC 160. The DAC 160converts the digital baseband transmit signal to an analog signal thatis supplied to the transmitter 110 over connection 134. The modulator116 and the upconverter 117 modulate and upconvert the analog transmitsignal according to the modulation format prescribed by the system inwhich the portable transceiver 100 is operating. The modulated andupconverted transmit signal is then supplied to the power amplifier 140over connection 112.

When receiving, the filtered and downconverted receive signal issupplied from the receiver 120 to the ADC 170 over connection 136. TheADC digitizes the analog receive signal and provides the analog basebandreceive signal to the baseband subsystem 130 over connection 138. Thebaseband subsystem 130 recovers the received information.

FIG. 2 is a schematic diagram of an embodiment of a known single-endedvoltage-mode downconverter implemented as a passive mixer using anapproximate 25% duty cycle topology. A passive mixer is an example of animplementation of the downconverter 200 of FIG. 1. Although voltage modeoperation is illustrated in the embodiment shown in FIG. 2, a currentmode implementation can also be used. FIG. 2 illustrates an example ofutilizing 25% duty cycle LO signals to control the mixer switching. Inpractice less than 25% duty cycle might be desirable to prevent overlapbetween the on-times of the switches.

In a voltage mode mixer implementation, such as shown in FIG. 2,reducing the duty cycle to 20% or below is possible, but it also quicklyreaches the point of diminishing returns where noise contributions dueto aliasing of undesired input signals or noise around harmonics of theLO frequency degrade performance. Duty cycle between 20-25% is chosen inthis implementation. In the topology shown in FIG. 2, LO and 2LOmultiplication (described in greater detail in FIG. 3) is done in the LOpath rather than the RF path.

The voltage signal on connection 146 is provided to switches 222, 224,226 and 228. The switches 222, 224, 226 and 228 can be implemented usingany switch technology such as, for example, bipolar junction transistor(BJT) technology, field effect transistor (FET) technology, or any otherswitching technology. The switches 222, 224, 226 and 228 can also beimplemented using pass gates, each of which are typically implemented bya combination of an NFET and PFET transistor, as known in the art. Theswitches 222, 224, 226 and 228 are illustrated in FIG. 2 as simplesingle-pole single-throw switches to illustrate that any type ofswitches can be used to generate the switching signals described herein.

In the embodiment described herein, the in-phase (I) andquadrature-phase (Q) signals are differential. Therefore, the I signalincludes a V_(I+) signal and a V_(I−) signal. Similarly, the Q signalincludes a V_(Q+) signal and a V_(Q−) signal. The switch 222 generatesthe I+ signal, the switch 224 generates the I− signal, the switch 226generates the Q+ signal and the switch 228 generates the Q− signal. Theclock signals that drive the switches 222, 224, 226 and 228 areillustrated as having a 25% duty cycle and can be generated as will bedescribed below. The clock signal 232 drives the switch 222, the clocksignal 234 drives the switch 226, the clock signal 236 drives the switch224 and the clock signal 238 drives the switch 228. In accordance withproviding an approximate 25% duty cycle topology, none of the clocksignals 232 through 238 have any time period during which they overlap,or which are positive at the same time.

The output of the switch 222 is terminated by a capacitance 256 and aresistance 257, and is provided to one input of the amplifier 252. Theoutput of the switch 224 is terminated by a capacitance 258 and aresistance 259, and is provided to the other input of the amplifier 252.The output of the switch 226 is terminated by a capacitance 266 and aresistance 267, and is provided to one input of the amplifier 262. Theoutput of the switch 228 is terminated by a capacitance 268 and aresistance 269, and is provided to the other input of the amplifier 262.The output of the amplifier 252 on connection 254 is the differentialV_(I+) and V_(I−) output signal; and the output of the amplifier 262 onconnection 264 is the differential V_(Q+) and V_(Q−) output signal.

FIG. 3 is a graphical illustration showing the LO signals utilized by anembodiment of the passive mixer 200 described in FIG. 2. The in-phase LOsignal includes differential components LO_I and LO_I. Thequadrature-phase LO signal includes differential components LO_Q andLO_Q. The 2LO signal is an LO signal that occurs at twice the frequencyof the I and Q LO signals. The inverse of the 2LO signal is referred toas 2LO.

The 2LO signal is shown at trace 302, the LO_I signal is shown at trace304, and the LO_I signal is shown as trace 305. The LO_Q signal is shownat trace 306 and the LO_Q signal is shown as trace 307. These fivesignals are combined as follows to generate the four LO waveforms thatare applied to the downconverter 200.

The 2LO*LO_I signal is shown at trace 308. The signal 308 represents theLO_I+ signal. The 2LO*LO_I signal is shown at trace 312. The signal 312represents the LO_I− signal. The 2LO *LO_Q signal is shown at trace 314.The signal 314 represents the LO_Q+ signal. The 2LO *LO_Q signal isshown at trace 316. The signal 316 represents the LO_Q− signal.

The effective in-phase differential LO signal, eLO_I, is shown as trace318 and the effective quadrature-phase differential LO signal, eLO_Q, isshown as trace 322. These signals are derived respectively asLO_I+−LO_I− and LO_Q+−LO_Q−. As shown in FIG. 3, the effective in-phasedifferential LO signal, eLO_I, 318 and the effective quadrature-phasedifferential LO signal, eLO_Q, 322 provide an approximate 25% duty cycleat each polarity and ensure that switching takes place only on thetransitions of the 2LO signal 302, thus minimizing any influence ofswitching noise, and minimizing any I and Q signal overlap due to theLO_I signal 304 and the LO_Q signal 306. The trace 326 is acontinuous-wave example showing the sampling of an RF input signal bythe I+ signal 328, the Q+ signal 332, the I− signal 334 and the Q−signal 336.

FIG. 4 is a schematic diagram illustrating an embodiment of a low noisereceiver 400. According to the 3GPP standard, the low noise receiver 400should be able to demodulate a desired signal at strength ofapproximately −99 dBm, in the presence of a 0 dBm out-of-bandnon-spurious blocker at greater than 20 MHz offset from the desiredreceive frequency, or in the presence of a −43 dBm out-of-band spuriousblocker, such as one that may occur at a harmonic of the desired receivefrequency.

The low noise receiver 400 receives a signal from an antenna 138 thatsupplies the received signal to a front end module 144. The front endmodule 144 comprises, in this example, an antenna filter 402 thatsupplies the filtered signal to a transmit receive (T/R) switch module404. In the embodiment shown in FIG. 4, the T/R switch module 404 is asingle pole four-throw (SPFT) switch that switches transmit high band,transmit low band (circuitry not shown for simplicity); and receive highband and receive low band. In this quad-band example, the transmitreceive switch module 404 can be implemented using any type of switchesas known in the art.

The receive signal is provided from the appropriate switch elementwithin the T/R switch module 404 to a low pass filter module 410. In theembodiment shown in FIG. 4, the low pass filter module 410 includescircuitry for both the receive low band and the receive high band. Thelow pass filter module 410 operates as a harmonic reject filter, and asan impedance matching network. The low pass filter module 410 attenuatesout of band blocking signals that may occur at an odd harmonic, forexample the third and fifth harmonic, of the desired receive frequency;and also provides impedance matching from the T/R switch module 404 tothe input of the downconverter 200. In an embodiment, the inductors 412and 417 can have a value of 10 nanohenrys (nH) and the capacitors 414and 416 can have a value of 3.0 picofarads (pF); and the inductors 418and 422 can have a value of 3.3 nH and the capacitors 419 and 421 canhave a value of 1.5 pF.

The low band filter circuitry comprises an inductor 412, a capacitor414, an inductor 417 and a capacitor 416. Similarly, the high bandfilter circuitry comprises an inductor 418, a capacitor 419, an inductor422 and a capacitor 421. In an embodiment, the low pass filter module410 provides impedance matching from the relatively low impedance sourceto the relatively high impedance load and in the process, provides avoltage gain by acting as a step-up transformer, as known in the art. Asan example, the input of the low pass module 410 has an impedance ofapproximately 50Ω, which should be matched to the approximate 400Ωimpedance at the input to the downconverter 200. A filter networkproviding such a match will step up the voltage by SQRT(400/50), whichin dB is 20*log (SQRT(400/50)) =9dB.

The low noise receiver 400 also includes an embodiment of thedownconverter 200 shown in FIG. 2. In the example shown in FIG. 4, thedownconverter 200 is a two band low noise passive mixer comprisingtransistor switches 424, 426, 427 and 428 for the low band andtransistor switches 429, 431, 432 and 434 for the high band. Only thehigh band or low band switches are employed at a time, according to theband of operation. According to this embodiment, the transistor switches424, 426, 427, and 428, or the transistor switches 429, 431, 432, and434, are switched according to a 25% local oscillator (LO) duty cycle,with the LO waveforms and their phases as described in FIGS. 2 and 3.According to this operation, no two transistor switches in either of thehigh band or low band segments of the downconverter 200 will beoperating at the same instant.

The 25% duty cycle LO drive for the transistor switches 424, 426, 427,and 428, or the transistor switches 429, 431, 432, and 434, providesisolation between the I and Q baseband outputs on the capacitors, C_(L)of FIG. 4, by connecting only one of the capacitors to the single-endedRF input at any given instant. This prevents charge sharing between Iand Q capacitors, enhancing mixer gain, noise figure (NF) and thequality factor (Q) of the band pass filtering response at the RF inputof the downconverter 200. Single-ended to differential conversion inthis voltage mode sample-and-hold topology has the advantage ofapproximately 6 dB of additional voltage gain. It can be shown that thegain in this topology approaches 5.1 dB due to the sample/hold mixeroperation and single-ended to differential downconversion. Additionalgain due to impedance step-up from approximately 50 ohm (Ω) toapproximately 400Ω in the low pass filter 410 enhances the total gain toapproximately 14.1 dB from antenna input to passive mixer output. It isnoteworthy that this mixer gain is achieved without any active stages orbias current in the signal path. It should also be noted that thisfront-end design may benefit greatly from future technology scaling, asperformance of the passive switches and mixer LO generation circuitryimproves at lower gate lengths.

The output of the downconverter 200 is supplied to aresistive/capacitive (RC) filter network 436. Specifically, the outputof the transistor 424 or 429 is supplied to resistor 437 and capacitor438. The output of transistor 426 or 431 is supplied to resistor 439 andcapacitor 441. The output of transistor 427 or transistor 432 issupplied to resistor 442 and capacitor 444, and the output of transistor428 or transistor 434 is supplied to resistor 446 and capacitor 447.

The following description will be made with particular reference to theoutput of the transistor 424 and the filter network comprising resistor437 and capacitor 438 and the output of the transistor 426 and thefilter network comprising resistor 439 and capacitor 441 as an exampleonly. The balance of the circuit performs in the same manner. Thecapacitor 438 performs a sample-and-hold function and performssingle-ended to differential conversion for the signal output from thetransistor 424. Each time the transistor 424 is conductive for theperiod of time corresponding to the 25% duty cycle described above, theoutput of transistor 424 is stored on capacitor 438 to provide thesample-and-hold function. Then, with example reference to the in-phasesignal, the differential conversion is performed by the capacitor 438and the capacitor 441. The capacitor 438 charges during the interval 328(FIG. 3) and the capacitor 441 charges during the interval 334 (FIG. 3).Then, these outputs are differenced, resulting in a 2× magnitude becausethe signals are of opposite polarity. As an example, the value of thecombined signals is approximately 6 dB.

The resistors 437 and 439 provide a common-mode voltage (Vcm) because anon-zero common-mode voltage is used in a differential system that usesa single supply voltage. The parallel combination of the capacitor 438,resistor 437 and the resistance through the transistor 424 forms an RClow pass filter. In an embodiment these element values are chosen toprovide an RC low pass filter bandwidth of +/−1 MHz. It is this low passfilter response that is reflected through the downconverter 200 thatcauses a 2 MHz wide RF band pass response to appear at the input to thedownconverter 200, as is illustrated in FIG. 5.

The output of the RC network 436 is then supplied to a high gaintrans-admittance amplifier 450. In this embodiment, the low noisereceiver comprises four instances of the high gain trans-admittanceamplifier 450. The high gain trans-admittance amplifier 450 includes acurrent source 452, a transistor 454 and a resistor 456 configured toreceive an output of the resistor 437 and capacitor 438. Similarly theoutput of the resistor 439 and capacitor 441 is supplied to a high gaintrans-admittance amplifier comprising current source 457, transistordevice 458 and resistor 459. Similarly, the output of the resistor 442and the capacitor 444 is supplied to a high gain trans-admittanceamplifier comprising current source 461, transistor 462 and resistor464. Finally, the output of the resistor 446 and the capacitor 447 issupplied to a high gain trans-admittance amplifier comprising currentsource 466, transistor 467 and resistor 468. In an embodiment, thedownconverter 200 and the high gain trans-admittance amplifier 450 canoperate from a 1.2V regulated supply.

The output of the high gain trans-admittance amplifier 450 is providedto an RC lowpass filter 470. The RC lowpass filter 470 comprisesresistor 471, capacitor 472 and resistor 474. The RC lowpass filter 470also comprises resistor 476, capacitor 477, and resistor 478.

The output of the RC lowpass filter 470 is provided to a filter 480,comprising amplifier 481 and related resistors (R1 and R2) andcapacitors (C1 and C2), and amplifier 491 and related resistors (R1 andR2) and capacitors (C1 and C2). The filters 470 and 480 are notcompletely independent and affect each other due to loading at theirinterface. The composite characteristics of the filters 470 and 480 canbe adjusted using resistors 471, 476, capacitors 472 and 477, resistorR1, resistor R2, capacitor C1 and capacitor C2 to obtain a desiredfilter response. The overall receiver gain can be scaled using resistors456, 459, 464 and 468 or adjusting resistors 471 and 476, capacitors 472and 477, resistor R1, resistor R2, capacitor C1 and capacitor C2. Theconcept is not limited to the use of the particular active filtertopology shown; other topologies may be used including otherop-amp-based active filter topologies as well as passive RC filters.

The output voltage of the filter 480 is provided to an analog-to-digitalconverter (ADC) 490. The output voltage of the amplifier 481 is providedto the ADC 492, and the output voltage of the amplifier 491 is providedto the ADC 494. The digital output of the ADC 490 is provided to thebaseband subsystem 130.

FIG. 5 is a graphical illustration 500 showing an example frequencyspectrum within which the low noise receiver operates. The abscissa 502represents frequency and the ordinate 504 represents signal level. Theregion 506 illustrates the receive frequency range from 925 MHz to 960MHz. The region 506 also illustrates the filter region that would beprovided by a SAW filter if a SAW filter were present in the system. Thesignal 508 represents the desired signal and the region 512 depicts a 2MHz wide frequency response covering region 518, centered at the desiredreceive frequency (tuning frequency 516) that is provided by theoperation of the downconverter 200. In an embodiment, the downconverter200 can be referred to as a “filtering mixer.”

An out-of-band blocking signal, also referred to as an out-of-bandinterfering signal, is depicted in FIG. 5 using reference numeral 522.In this example, the out-of-band blocking signal 522 is approximately 20MHz higher in frequency than the upper frequency range of 960 MHz. Thedownconverter 200 exhibits the frequency response 512, thereby passingsignals within frequency range 518, and substantially rejecting signalsoutside frequency range 518, thereby preventing out-of-band blockingsignals from interfering with the desired signal 508. The frequencyresponse 512 is a band pass response with very high Q around the tuningfrequency 516 (the frequency of the LO (f_(LO))) having a 3 dB bandwidthof 2 MHz centered at the tuning frequency 516. This high Q band passresponse is established by the low pass pole due to capacitor 438 andresistor 437 of FIG. 4 (for example, C_(L) and R_(B)) being effectivelyreflected through the transistors in the downconverter 200 to present aband pass pole centered on the LO frequency at the downconverter input.For higher offsets around LO, a 20 dB/decade drop in input impedance isobserved, until the response reaches a floor that is determined by thefinite on resistance of the passive switches used in the downconverter200. By means of this high Q filter at the downconverter input, a 20 MHzblocker in the GSM 950 MHz band is attenuated by more than 12 dB.

As the local oscillator frequency applied to the downconverter 200 ofFIG. 4 changes, the 2 MHz wide region 512 will shift with the tuningfrequency 516. Any channel to which the receiver 400 is tuned will havethis 2 MHz wide filter region around the tuning frequency 516, thuseliminating any out-of-band (beyond 2 MHz) blocking signals. Thiseliminates the need for the SAW filter at the input to the low noisereceiver 400.

This ‘tracking filter” operation together with the low noise provided bythe downconverter 200 allows the elimination of a low noise amplifier,as shown in FIG. 4 where the front end module 144 is connected directlyto the low pass filter 410 at the input to the downconverter 200. The25% duty cycle LO, derived by the LO 2LO method described in FIG. 3,applied to the downconverter 200, providing non-overlappingdownconverter phases as shown in FIG. 3, allows approximately 6 dBvoltage gain to be provided by the downconverter 200, thus furtherjustifying the omission of a low noise amplifier between the front endmodule 144 and the low pass filter 410.

However, if the out-of-band blocking signal 522 occurs at a frequencythat is either three or five times the tuning frequency 516 of thedesired signal 508 (commonly referred to as the third or fifth harmonicof the fundamental frequency), then, through a phenomenon referred to asmixer aliasing, the full amplitude of the out-of-band blocking signal522 would be superimposed over the desired signal 508, thus degradingreceiver sensitivity at the tuning frequency 516.

In order to prevent an out-of-band blocking signal 522 that may occur atan odd harmonic, for example, the third or fifth harmonic, of thedesired signal 508 from interfering with the desired signal 508, the lowpass filter 410 (FIG. 4) is implemented to reduce receiver sensitivityat the third and fifth harmonic frequency of the desired signal 508. Thetotal number of matching components used in the low pass filter 410 isless than or equal to that used in typical quad-band receiver matchingcircuits. A simple fourth order filter provides more than 30 dBrejection for the unwanted components at three or five times the desiredreceive frequency. By appropriate choice of components, this rejectioncan be increased to more than 65 dB by the utilization of componentself-resonances.

Further, as will be described below in FIG. 7, taking advantage of theoutput phases available from the downconverter 200, the phases can besummed in order to further attenuate out-of-band blocking signalsoccurring predominately at odd harmonics, for example, at the third andfifth harmonics, of the desired signal.

FIG. 6 is a schematic diagram illustrating an alternative embodiment ofthe low noise receiver of FIG. 4. Elements in FIG. 6 that are similar toelements in FIG. 4 will be numbered using the convention 6XX, where the“XX” in FIG. 6 refers to a similar element in FIG. 4. Further, some ofthe reference numerals in FIG. 6 are not shown for simplicity. The lownoise receiver 600 is similar to the low noise receiver 400, except thatthe embodiment of FIG. 6 shows an exemplary baseband filterimplementation where the output current from the baseband V-I conversionstage provided by a high gain trans-admittance amplifier 650 is applieddirectly to the virtual ground of a continuous time ADC 690, comprisingADC elements 692 and 694, after passive low pass filtering in the RClowpass filter 670.

FIG. 7 is a schematic diagram illustrating another alternativeembodiment of the low noise receiver of FIG. 4. The embodiment of thelow noise receiver of FIG. 7 illustrates only one band (the low band)and shows an example of generating eight (8) output phases of thedownconverter 200. Additional attenuation of out-of-band blockingsignals that may occur at odd harmonics, for example at the third andfifth harmonics of the desired receive frequency, can be obtained bytaking advantage of the output phases available from the downconverter200. The output phases from the downconverter 200 can be summed in orderto further attenuate out-of-band blocking signals at, for example, thethird and fifth harmonics of the desired signal.

The embodiment of the low noise receiver 700 illustrates only the lowband for simplicity of illustration. The low noise receiver 700 includesan implementation of a downconverter 200 shown using simple switchesinstead of transistor devices and illustrates only the low band (LB)signal chain for simplicity. The LO drive signals for the switches areshown using the graphical illustration 750. The embodiment of thedownconverter 715 includes 2K taps, taking a total of 2K samples percomplete cycle of the LO frequency. In a general 2K tap downconverter715, the duty cycle of each LO waveform is less than LO/2K. The gain ofthe downconverter 715 approaches 0 dB as K increases. For the case of asingle-ended downconverter, the gain approaches 6 dB from thecombination of single-ended to differential conversion and the sampleand hold (S/H) operation described above. Any voltage step-up in the lowpass filter module 710 provides additional gain, as discussed above.

The 2K tap implementation where K is 4, 8, 16, etc., allowsconfigurations where harmonics of the input RF frequency can be rejectedby simple weighted summation of the outputs of the downconverter 715. Anexample of the summation of three output phases that provide a waveformthat carries no third or fifth harmonics is described in FIG. 8.

The signal from the low pass filter module 710 is provided todownconverter 715 which is shown for simplicity as an array of switches.Each switch is shown with the designation of the LO waveform 750 thatdrives it (LO_0 through LO_(2K−1)). In the general implementation shownin FIG. 7, 2K switches (LO_0 through LO_(2K−1)) are used in the signalpath, each switch having a duty cycle≤(100/2K) %. The period of the LOfrequency is T, and each LO waveform exhibits an active pulse width ofT/2K. The implementation discussed in this example is a specific casefor K=4, so each LO waveform 750 exhibits an active pulse width of T/8.However, any number K of baseband outputs could be used in receivertopologies depending on application. As the number K increases, thesample-and-hold gain approaches 0 dB. For example, a 3^(rd) and 5^(th)harmonic rejection receiver architecture might use K=4 to generate the0, 45, 90, 135, 180, 225, 270 and 315 degree samples of the RF waveform.The outputs denoted by V(0), V(1), . . . V(2K−1) in FIG. 7, for the caseof K=4, correspond to the 0, 45, 90, 135, 180, 225, 270 and 315 degreesamples, respectively. The outputs V(0), V(1), . . . V(2K−1) are groupedin pairs where each pair comprises outputs differing in phase by 180degrees. For instance, the difference of V(0) and V(K), the differenceof V(1) and V(K+1), and the difference of V(K−1) and V(2K−1). Thedifference of each of these pairs is then determined by a respectivedifference amplifier 785-1 through 785-K. Difference amplifiers 785-1through 785-K may also include low pass filters, as described in FIG. 4as filters 480. Since the signals being differenced are 180 degrees outof phase, 6 dB gain is achieved. In the specific case for K=4, theresulting outputs of difference amplifiers 785-1 through 785-K representgain-added phases of the received signal at 0, 45, 90, and 135 degreeswith the added 6 dB gain. The outputs of difference amplifiers 785-1through 785-K are applied to ADCs 790-1 through 790-K. Outputs of ADCs790-1 through 790-K are then applied to baseband system 130. Withinbaseband system 130, harmonic rejection summing can be implemented usingweighted summations of these multiple phases, as will be describedbelow.

The technique shown in FIG. 7 is an effective way of splitting the RFsignal in the time domain into K separate paths without adding extracircuit blocks that could severely degrade performance or increase powerconsumption and die area.

FIG. 8 illustrates a known method of generating a waveform in which3^(rd) and 5^(th) harmonics are rejected. FIG. 8 shows only the signalsrelating to the in-phase (I) signal. For simplicity, the example in FIG.8 shows an example of the summation of three output phases that providea waveform that carries no third or fifth harmonics. Other numbers ofoutput phases can be combined to achieve a similar output waveform.

The waveform 820 represents the fundamental LO signal according to theequation:

${U\; 1(t)} = {\frac{2}{\pi}\left\lbrack \left( {{\cos\;\left( {\omega\; t} \right)} - {{1/3}\left( {{\cos\left( {3\omega\; t} \right)} + {{1/5}\left( {{\cos\left( {5\omega\; t} \right)}\mspace{14mu}\ldots}\mspace{14mu} \right\rbrack}} \right.}} \right. \right.}$

The waveform 810 represents the fundamental LO signal 820 advanced 45degrees relative to the signal 820. Signal 810 is represented accordingto the equation:

${U\; 2\;(t)} = {\frac{\sqrt{2}}{\pi}\left\lbrack {\left( {{\cos\;\left( {\omega\; t} \right)} - {\sin\;\left( {\omega\; t} \right)}} \right) + {{1/3}\left( {{\cos\left( {3\omega\; t} \right)} + {\sin\;\left( {3\omega\; t} \right)}} \right)} - {{1/5}\left( {{\cos\left( {5\omega\; t} \right)} - {\sin\;\left( {5\omega\; t} \right)}} \right)\mspace{14mu}\ldots}}\mspace{14mu} \right\rbrack}$

The waveform 830 represents the fundamental LO signal 820 retarded by 45degrees relative to signal 820. Signal 830 is represented according tothe equation:

${U\; 3(t)} = {\frac{\sqrt{2}}{\pi}\left\lbrack {\left( {{\cos\;\left( {\omega\; t} \right)} + {\sin\;\left( {\omega\; t} \right)}} \right) + {{1/3}\left( {{\cos\left( {3\omega\; t} \right)} - {\sin\;\left( {3\omega\; t} \right)}} \right)} - {{1/5}\left( {{\cos\left( {5\omega\; t} \right)} + {\sin\;\left( {5\omega\; t} \right)}} \right)\mspace{14mu}\ldots}}\mspace{14mu} \right\rbrack}$

The waveform 840 represents the combination of the above three waveformsin the appropriate proportions such that the third and fifth harmonicsof the fundamental LO signal 820 are rejected. The combination is formedaccording to the equation:LO_harm_rej(t)=√{square root over (2)}U1(t)+U2(t)+U3(t)

Returning now to FIG. 7, the effective LO outputs of the downconverter715 can be combined as generally described above with respect to FIG. 8,and as will be described below in FIG. 10 for the case of eight outputphases, to provide additional harmonic rejection, which furthersimplifies the requirements for the low pass filter 410 (FIG. 4). ForK=4, a downconverter configuration is obtained that provides third andfifth harmonic rejection, allowing the receiver to reject input signalsat three times and five times the desired RF signal.

FIG. 9 is a graphical illustration showing the eight LO phases utilizedby the low noise receiver of FIG. 7 for the case of K=4. The trace 902shows a 4LO waveform with 50% duty cycle. The traces 904 and 906 showtwo quadrature phases of 2LO, respectively referred to as 2LO_I and2LO_Q. The traces 908 and 912 show two 45 degree offset phases of LO,respectively referred to as LO_I and LO_Q. The signals represented bythe traces 902, 904, 906, 908 and 912 are multiplied in the eightcombinations shown by traces 922, 924, 926, 928, 932, 934, 936 and 938to produce eight respective LO waveforms, referred to as LO_0, LO_4,LO_1, LO_5, LO_2, LO_6, LO_3 and LO_7, each of which exhibits a ⅛ dutycycle.

FIG. 10 is a graphical illustration showing the effective quadrature LOwaveforms, each with 3^(rd) and 5^(th) harmonics rejected, that aregenerated by weighted combining of the eight LO phases utilized by thelow noise receiver of FIG. 7 for the case of k=4.

In FIG. 10, the eight ⅛-duty-cycle waveforms LO_0 through LO_7, shownrespectively by traces 922, 924, 926, 928, 932, 934, 936 and 938 arecombined in the baseband subsystem 130 in the appropriate proportions toform effective quadrature waveforms eLO_I 1002 and eLO_Q 1004, as willbe further illustrated in FIG. 11 and FIG. 13. The waveforms eLO_I 1002and eLO_Q 1004 exhibit the same harmonic-rejecting characteristic shapefor a signal having eight combined output phases as that shown by trace840 in FIG. 8 for a combination of three output phases.

Suppression of harmonics greater than the fifth harmonic can be achievedby increasing the number of output phases. For example, using 16 outputphases and the proper choice of weighting coefficients, a frequencyresponse suppressing the 3rd, 5th, 7th, 9th, 11th, and 13th, harmonicscould be achieved. Such a response would look similar to the plot 1220(FIG. 12), extended out to 16 GHz, with big lobes only at 1 GHz and 15GHz. In such a case the waveforms eLO_I and eLO_Q would exhibit afiner-toothed quantization compared to the plots 1002 and 1004 in FIG.10. As the number of output phases further increases toward infinity,eLO_I and eLO_Q would become pure sine waves, which contain no harmonicsat all.

FIG. 11 is a schematic diagram illustrating an embodiment of a low noisereceiver that implements the effective quadrature LO waveforms of FIG.10. The low noise receiver 1100 is an alternative embodiment of the lownoise receiver 700 of FIG. 7 and combines the effective quadrature LOwaveforms of FIG. 10 to provide additional rejection of 3rd and 5^(th)harmonics at the input of the downconverter. The switches that comprisethe downconverter 1115 are controlled by the 8 LO phases shown by traces922, 924, 926, 928, 932, 934, 936 and 938 in FIG. 9. The embodimentshown in FIG. 11 includes 8 phases of the LO signal, and as such, the 8LO signals are represented as LO_0 through LO_7, as shown in thegraphical illustration 1150.

Combining the eight LO phases to provide additional rejection of the3^(rd) and 5^(th) harmonics occurs in two parts. The first combining ofthe 8 LO phases occurs in the analog domain using analog differenceamplifiers 1185-1, 1185-2, 1185-3 and 1185-4. Every nth sample of thereceived signal is differenced with the (n+4)th sample by the respectiveanalog difference amplifiers 1185. The LO_0 signal is combined with theLO_4 signal by the analog difference amplifier 1185-1. The LO_1 signalis combined with the LO_5 signal by the analog difference amplifier1185-2. The LO_2 signal is combined with the LO_6 signal by the analogdifference amplifier 1185-3. The LO_3 signal is combined with the LO_7signal by the analog difference amplifier 1185-4. The respective outputsof the analog difference amplifiers 1185-1 through 1185-4 representphases of the received signal at 0, 45, 90 and 135 degrees with anapproximate 6 dB added gain, as described above in FIG. 7.

The outputs of the analog difference amplifiers 1185 are converted tothe digital domain by respective ADC elements 1190. The output of theanalog difference amplifier 1185-1 is supplied to the ADC 1190-1. Theoutput of the analog difference amplifier 1185-2 is supplied to the ADC1190-2. The output of the analog difference amplifier 1185-3 is suppliedto the ADC 1190-3. The output of the analog difference amplifier 1185-4is supplied to the ADC 1190-4.

The second combining of the eight LO phases occurs in the digital domainusing a digital summation harmonic reject filter 1125, which can beimplemented in hardware, software, or a combination of hardware andsoftware. In an embodiment, the digital summation harmonic reject filter1125 is part of the operation of the receiver software 155 and isexecuted by the processor 135. The receiver software 155 performs asummation represented by summation elements 1130 and 1132. The output ofthe ADC 1190-1 is provided to multiplying element 1142 and to themultiplying element 1144. The output of the ADC 1190-2 is provided tomultiplying element 1146 and to the multiplying element 1148. The outputof the ADC 1190-3 is provided to multiplying element 1152 and to themultiplying element 1154. The output of the ADC 1190-4 is provided tomultiplying element 1156 and to the multiplying element 1158. Eachmultiplying element digitally amplifies the signal passing though it byits respective weighting factor shown in FIG. 11. For example, theoutput of ADC 1190-1 is digitally amplified by multiplying element 1142by a factor of 1+√{square root over (2)}/2. The summation of theweighted signals is performed in the summation elements 1130 and 1132,resulting in baseband outputs I and Q. Importantly, the switches in thedownconverter 1115 do not interfere with one another due to thenon-overlapping LO signals that drive them. Further, the summationperformed by the summation elements 1130 and 1132 is done at baseband,but has the effect of rejecting harmonics, particularly 3^(rd) and5^(th) harmonics, at RF. Therefore the baseband outputs I and Qrepresent a faithful reproduction of the baseband signals carried on thedesired RF carrier to which the receiver is tuned, without anysubstantial interference due to the presence of undesired RF blockingsignals that may exist at the 3^(rd) and 5^(th) harmonics of the desiredRF carrier.

FIG. 12 is a graphical illustration showing an example of the frequencyresponse of embodiments of the low noise receiver. The example in FIG.12 shows the response for a 1 GHz receive signal. The plot 1210illustrates the effective response of the switching and summing actionsof downconverter 200 of FIG. 4, for which K=2. Plot 1210 does notinclude the effect of LC harmonic rejection filter 410. In plot 1210 theeven harmonics are rejected but the odd harmonics remain. Thus, the LCharmonic rejection filter 410 in FIG. 4 must provide all the attenuationof the 3^(rd), 5^(th), and 7^(th) harmonics.

The plot 1220 illustrates the effective response of the switching andsumming actions of the low noise receiver 1100 of FIG. 11, for whichK=4. Plot 1220 does not include the effect of any LC antenna filter. Inthis case, the 3^(rd) and 5^(th) harmonics are greatly rejected due tothe 8-phase switching and the harmonic-rejection summation, leaving onlythe 7^(th) harmonic. Thus, when an LC antenna filter is added to thesystem of FIG. 11 such LC antenna filter need only reject the 7^(th)harmonic, which is far easier than rejecting the 3^(rd) and 5^(th)harmonics as described above.

FIG. 13 is a schematic diagram illustrating an alternative embodiment ofthe low noise receiver of FIG. 11. The embodiment 1300 shown in FIG. 13shows analog summation of the LO signals, LO_0 through LO_7 described inFIG. 10 for the case of K=4. The switches that comprise thedownconverter 1315 are controlled by the 8 LO phases shown by traces922, 924, 926, 928, 932, 934, 936 and 938 in FIG. 9. The embodimentshown in FIG. 13 includes 8 phases of the LO signal, and as such, the 8LO signals are represented as LO_0 through LO_7, as shown in thegraphical illustration 1350.

Combining the eight LO phases occurs in two parts. The first combiningof the 8 LO phases occurs in the analog domain using analog differenceamplifiers 1385-1, 1385-2, 1385-3 and 1385-4. Every nth sample of thereceived signal is differenced with the (n+4)th sample by the analogdifference amplifiers 1385. The LO_0 signal is combined with the LO_4signal by the analog difference amplifier 1385-1. The LO_1 signal iscombined with the LO_5 signal by the analog difference amplifier 1385-2.The LO_2 signal is combined with the LO_6 signal by the analogdifference amplifier 1385-3. The LO_3 signal is combined with the LO_7signal by the analog difference amplifier 1385-4.

In this embodiment, the second combining of the eight LO phases alsooccurs in the analog domain in an analog summation reject filter 1325.The filter 1325 performs a summation using summation elements 1330 and1332. The output of the analog difference amplifier 1385-1 is providedto amplifier 1342 and to the amplifier 1344. The output of the analogdifference amplifier 1385-2 is provided to amplifier 1346 and to theamplifier 1348. The output of the analog difference amplifier 1385-3 isprovided to amplifier 1352 and to the amplifier 1354. The output of theanalog difference amplifier 1385-4 is provided to amplifier 1356 and tothe amplifier 1358. Each amplifier 1342, 1344, 1346, 1348, 1352, 1354,1356 and 1358 amplifies the signal passing though it by its respectiveweighting factor shown in FIG. 13. For example, the output of analogdifference amplifier 1385-1 is amplified by amplifier 1342 by a factorof 1+√{square root over (2)}/2. The summation of the weighted signals isperformed in the summation elements 1330 and 1332, resulting in analog Iand Q signals. Importantly, the switches in the downconverter 1315 donot interfere with one another due to the non-overlapping LO signalsthat drive them.

Further, the summation performed by the summation elements 1330 and 1332is done at baseband, but has the effect of rejection of harmonics at RF.Therefore the baseband outputs I and Q represent a faithful reproductionof the baseband signals carried on the desired RF carrier to which thereceiver is tuned, without any substantial interference due to thepresence of undesired RF blocking signals that may exist at the 3^(rd)and 5^(th) harmonics of said desired RF carrier.

The in-phase output of the summing element 1330 is provided to an ADC1395 for conversion to the digital domain. The quadrature-phase outputof the summing element 1332 is provided to an ADC 1396 for conversion tothe digital domain. The digital in-phase signal and the digitalquadrature-phase signal are then provided to the baseband subsystem 130(FIG. 1) for further processing.

Typical rejection for harmonics of the desired signal frequency, withharmonic rejection summation performed in the analog domain as shown inFIG. 13, is limited to approximately 35 dB to 40 dB, due to analogcomponent tolerances, while the digital implementation shown in FIG. andFIG. 11 can achieve greater than 40 dB rejection since the only analogtolerances remaining in the implementation of FIG. 11 are those of thesampling capacitors, difference amplifiers and ADCs. Proportionalsummation in the digital domain, as shown in FIG. 11 potentially allowsfor implementation of least mean squares (LMS) based algorithms that canmaximize the rejection at n times the desired signal by furthercompensating for any analog mismatches in the various paths.

FIGS. 14A through 14D are graphical illustrations showing an examplefrequency response of an embodiment of the low noise receiver of FIG. 11or FIG. 13, with the addition of a low pass filter module between theantenna and the input to the downconverter, operating with a receivefrequency of 1 GHz. FIG. 14A illustrates an example response of a4^(th)-order low pass filter module 410. In this example the filter isdesigned with a wide bandwidth and gentle slope, as it is required toprovide rejection only at the 7^(th) harmonic, and not the 3^(rd) or5^(th) harmonic. FIG. 14B shows the 2 MHz-wide passband at 1 GHz (causedby the switching and the RC), plus all the unwanted similar responsesthat occur at harmonics due to aliasing. FIG. 14C shows the responseformed by the harmonic-rejection summation of FIG. 11 or FIG. 13. FIG.14D shows the cascaded response of FIGS. 14A, 14B and 14C. In thecascaded response of FIG. 14D is shown the desirable characteristics ofthe 2 MHz-wide response that tracks the receiver's tuned frequency, withsimilar responses rejected at the 3^(rd) and 5^(th) harmonics, andsubstantially suppressed at the 7^(th) harmonic.

While various embodiments of the invention have been described, it willbe apparent to those of ordinary skill in the art that many moreembodiments and implementations are possible that are within the scopeof the invention. For example, the invention is not limited to aspecific type of radio receiver or transceiver. Embodiments of theinvention are applicable to different types of radio receivers andtransceivers and are applicable to any receiver that downconverts orfilters a received signal.

What is claimed is:
 1. A wireless communication device comprising: aradio frequency antenna; and a transceiver including a basebandsubsystem and a receiver, the receiver coupled between the antenna andthe baseband subsystem and including a filter, the filter configured toimplement low pass filtering and gain functions on a radio frequencysignal and to output a filtered radio frequency signal, the receiverfurther including a downconverter configured to generate a plurality ofoutput phases within a local oscillator period based on (i) the filteredradio frequency signal and (ii) a respective plurality of localoscillator signals, the plurality of output phases organized into atleast K groups where K is an integer of four or greater, each nth groupof the K groups including nth and (n+K)th output phases of the pluralityof output phases, n being an integer between 0 and K−1, and the receiverfurther including a differencing stage configured to difference the nthand (n+K)th output phases of each respective group of the K groups,resulting in gain-added output phases, the filter further configured toprovide impedance matching at an input of the downconverter, theimpedance matching providing a step up in voltage by a factor of asquare root of the ratio of an input impedance of the downconverter toan input impedance at an input to the filter.
 2. The wirelesscommunication device of claim 1 wherein the downconverter provides again of at least approximately 6 decibels.
 3. The wireless communicationdevice of claim 1 wherein the receiver does not include a surfaceacoustic wave filter.
 4. The wireless communication device of claim 3wherein the receiver does not include a low noise amplifier foramplifying the radio frequency signal.
 5. The wireless communicationdevice of claim 1 wherein K is a power of
 2. 6. The wirelesscommunication device of claim 1 wherein K is 4, 8, or
 16. 7. Thewireless communication device of claim 1 wherein K is a multiple of 4.8. A radio frequency receiver comprising: a filter configured to lowpass filter a radio frequency signal and output a filtered radiofrequency signal; a downconverter configured to generate a plurality ofoutput phases within a local oscillator period based on the filteredradio frequency signal, the plurality of output phases organized into atleast K groups where K is an integer of four or greater, where each nthgroup of the K groups includes nth and (n+K)th output phases of theplurality of output phases, and where n is an integer between 0 and K−1;and a differencing circuit configured to difference the nth and (n+K)thoutput phases of each respective group of the K groups, resulting ingain-added output phases, the filter configured to provide impedancematching at an input of the downconverter, the impedance matchingproviding a step up in voltage by a factor of a square root of the ratioof an input impedance of the downconverter to an input impedance at aninput to the filter.
 9. The radio frequency receiver of claim 8 whereinK is a power of
 2. 10. The radio frequency receiver of claim 8 wherein Kis 4, 8, or
 16. 11. The radio frequency receiver of claim 8 wherein K isa multiple of four.
 12. The radio frequency receiver of claim 8 whereinthe plurality of output phases includes 8 or 16 output phases.
 13. Theradio frequency receiver of claim 8 wherein the downconverter provides again of at least approximately 6 decibels.
 14. The radio frequencyreceiver of claim 8 wherein the receiver does not include a surfaceacoustic wave filter or a low noise amplifier for amplifying the radiofrequency signal.
 15. A wireless communication device comprising: anantenna configured to wirelessly transmit and receive radio frequencysignals; and a receiver in communication with the antenna including afilter and a downconverter, the filter configured to output a filteredradio frequency signal, the downconverter configured to generate aplurality of output phases within a local oscillator period based on thefiltered radio frequency signal, the plurality of output phasesorganized into at least K groups where K is an integer of four orgreater, each nth group of the K groups includes nth and (n+K)th outputphases of the plurality of output phases, where n is an integer between0 and K−1, the receiver further including a differencing stageconfigured to difference the nth and (n+K)th output phases of eachrespective group of the K groups, resulting in gain-added output phasesof the filtered radio frequency signal, the filter configured to provideimpedance matching at an input of the downconverter, the impedancematching providing a step up in voltage by a factor of a square root ofthe ratio of an input impedance of the downconverter to an inputimpedance at an input to the filter.
 16. The wireless communicationdevice of claim 15 wherein K is a power of
 2. 17. The wirelesscommunication device of claim 15 wherein K is 4, 8, or
 16. 18. Thewireless communication device of claim 15 wherein the plurality ofoutput phases includes 8 or 16 output phases.
 19. The wirelesscommunication device of claim 15 wherein the downconverter provides again of at least approximately 6 decibels.
 20. The wirelesscommunication device of claim 15 wherein the receiver does not include asurface acoustic wave filter or a low noise amplifier for amplifying theradio frequency signal.